Control of dynamic voltage restorer under voltage sag and nonlinear load
Nguyen Trong Huan, Ho Nhut Minh, Van Tan Luong
CONTROL OF DYNAMIC VOLTAGE
RESTORER UNDER VOLTAGE SAG
AND NONLINEAR LOAD
Nguyen Trong Huan*, Ho Nhut Minh*, Van Tan Luong+
* Học Viện Cô ng Nghệ Bưu Chính Viễn Thông Cơ Sở Thành Phố Hồ Chí Minh
+ Trường Đại học Cô ng nghiệp Thực phẩm Thành Phố Hồ Chí Minh
1Abstract - In this paper, a nonlinear control scheme for
dynamic voltage restorer (DVR) is proposed to reduce the
voltage disturbances for loads under grid voltage sags and
nonlinear loads. First, the nonlinear model of the system
consisting of LC filter is obtained in the dq0 synchronous
reference frame. Then, the controller design is performed
by using the sliding mode control, where the load voltages
are kept almost sinusoidal by controlling the dq0 axis
components of the DVR output voltages. With this
scheme, the power quality is significantly improved,
compared with the conventional proportional-integral (PI)
controller under grid voltage sags and nonlinear loads.
Simulation studies are performed to verify the validity of
the proposed method.
Conventionally, a cascaded controller including an
outer voltage control and inner current control loops has
been suggested [8]. However, its control dynamic response
is slow since the voltage control loop has the limitted
bandwidth [5]. Also, when there are unbalanced voltage
sags, the source voltage contains the negative sequence and
zero-sequence components and hence, the d-q components
of the source voltage can not be DC signals. Normally, a
typical PI (proportional integral) controller does not work
well for controlling the AC signals. Thus, a resonant
control scheme has been employed to regulate the unified
power quality conditioner, to compensate the load voltages
under unbalanced and distorted conditions of source
voltage and load [9]. Another issue considered for
controlling the UPS (uninterruptible power supply) or DVR
is the nonlinearity of the UPS or DVR [10], [11]. Thus, the
nonlinear control gives better performance than the control
techniques based on the PI control.
Keywords - Dynamic voltage restorer, nonlinear load,
sliding mode control, voltage sags.
In the paper, a control method based on a sliding mode
(SM) has been applied to improve the operation of the
three-phase four-wire (3P4W) DVR system under grid
fault conditions and nonlinear loads. First, the nonlinear
model of the system including LC filter is obtained in the
dq0 synchronous reference frame. Then, the controller
design depending on the sliding mode control is performed,
in which the load voltages are kept almost sinusoidal. The
simulation results show the validity of the proposed control
method.
I. INTRODUCTION
In recent years, as the penetration of the renewable
energy systems into the grid at the point of common
coupling (PCC) increases rapidly, the issues of the power
quality are paid much attention. The critical power quality
issues in distribution systems are related to grid voltage
disturbances. Since the application of power electronics
devices has been increased in industrial processes,
disturbances of the power supply affect the industrial loads.
This can cause malfunctions, tripping, or even faults of the
load system. The voltage sags, swells, harmonics,
unbalances, and flickers, known as power quality issues,
are generally considered as critical phenomena of voltage
disturbances in distribution systems, in which the voltage
sags is a main reason of short-circuit faults [1]-[4].
II. OVERVIEW OF DVR SYSTEM
A. System modeling
The three-phase DVR circuit in Figure 1 can be
represented in synchronous dq0 reference frame. Due to
conditions of grid voltage sags and nonlinear loads, the
dq0-axis components are taken into account as [11], [12]:
Several methods have been used to improve the power
quality in the distribution networks. A dynamic voltage
restorer (DVR) system is one of the best solutions which
keep the load voltage at its rated value when the grid
voltage drops occur suddenly. The DVR system is
composed of a voltage-source inverter (VSI), output LC
filters, and an isolated transformer connected between the
source and the loads [5]-[7]. Normally, both primary and
secondary coils of the transformer are connected in Y-
windings in distribution systems.
1
1
ifdq
vdq − vcdq − jifdq
(1)
(2)
Lf
Lf
1
1
if 0
=
v0 −
vc0
L +3L
L +3L
f
n
f
n
----------------------------------------------
Tác giả liên hệ: Nguyen Trong Huan
Email: huannt@ptithcm.edu.vn
Đến tòa soạn: 9/2020; chỉnh sửa: 11/2020; chấp nhận đăng: 12/2020.
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CONTROL OF DYNAMIC VOLTAGE RESTORER UNDER VOLTAGE SAG AND NONLINEAR LOAD
(vd*vr,dq0
1
1
voltage references
for the DVR system in the
)
vcdq
=
dq − ifdq − jvcdq
(3)
(4)
Cf
Cf
synchronous reference frame are calculated as:
vd*vr,dq0 =es,dq0 −vL*,dq0
(7)
1
1
vc0
f 0
−
i0
Cf
Cf
where
is the dq0-axis components of the source
es,dq0
voltage, and vL*,dq0 is the dq0-axis components of the load
where Lf, Ln, and Cf are the filter inductance, the neutral
filter inductance, and the filter capacitance, respectively;
vcdq0 are the dq0-axis capacitor voltages; vdq0 are the dq0-
axis inverter terminal voltages; idq0 are the dq0-axis output
currents of the DVR; ifdq0 are the dq0-axis output currents;
ω is the source angular frequency.
*
*
voltage references, in which both
and L,0 are also set
v
vL,d
to be zero and vL*,q is set to be magnitude of the load
voltage at the rating (vL,mag
)
.
From (1) to (4), a state-space modeling of the system is
derived as follows:
III. PROPOSED CONTROL STRATEGY USING
SLIDING MODE CONTROL
0
0
0
0
−1/ Lf
0
0
0
0
i
fd
i
A
multi-input multi-output (MIMO) nonlinear
fd
−
−1/ Lf
ifq
i
ifq
approach is proposed for the purpose of eliminating the
nonlinearity in the modeled system [10]. Consider a MIMO
system as follows:
1
0
0
0
0
0
−
if 0
f 0
(5)
Lf + 3Ln
=
v
v
cd
1/ Cf
0
1/ Cf
0
0
0
0
−
0
0
0
0
0
vcq
v
0
0
(8)
(9)
x f x+ g u
y = h(x)
vc0
vc
1/ Cf
0
1/ L
0
0
0
1
f
0
0
0
0
1/ Lf
where x is state vector, u is control input, y is output, f and
g are smooth vector fields, h is smooth scalar function.
0
0
v
d
Lf + 3Ln
+
+
id / Cf
i / Cf
vq
0
0
0
0
0
0
0
0
0
q
The dynamic model of the inverter in (5) is expressed in
(8) and (9) as:
i / C
0
f
T
x = ifd ifq if 0 vcd vcq vc0
;
Series Transformer
esa
esb
esc
isa
isb
isc
iLa
iLb
iLc
T
u = vd vq v0
;
Linear load and
Nonlinear load
T
y = vcd vcq vc0
To generate an explicit relationship between the outputs
and the inputs , each output is differentiated
S1
S3
S5
C1
yi=1,2,3
until a control input appears.
ui=1,2,3
ic
ib
ia
Lf
ifa
ifb
ifc
Vdc
C2
vca vcb vcc
y
u
1
1
Cf
S4
S6
S2
(10)
2 = x + E x u
( )
( )
2
L0
u3
Then, the control law is given as:
Figure 1. Circuit configuration of three-phase four-wire
DVR.
*
d
v
u
v
1
1
v* = u = E−1(x) −A(x) + v2
B. Generation of voltage references
(11)
q
2
In this research, the strategy of in-phase compensation
is considered, in which the amplitude of the load voltage is
exactly kept the same as before the sag, while the phase of
the load voltage is similar to that of the source voltage after
the sag. As shown in Figure 1, the load voltage is expressed
as:
v0*
v3
u3
where
2
1
1 1
id − q
Cf Cf
ifq
−
+2
v
−
cd
Cf
Lf Cf
vL,abc = es,abc − vdvr,abc
(6)
2
1
1
1
iq d
Cf
A x =
( )
−
−
ifd
−
+2 v −
cq
Cf
Lf Cf
Cf
where vL,abc is the load voltage, es,abc is the d-q axis capacitor
voltage, and vdvr,abc is the voltage injected by the DVR.
1
1
vc0
−
if 0
Cf
L +3L C
The control of the DVR is performed in the synchronous
reference frame, in which the phase angle of the source
voltage is used for transforming the DVR output voltages
and load voltages. To keep the load voltage constant, the
f
n
f
;
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Nguyen Trong Huan, Ho Nhut Minh, Van Tan Luong
V =
1
1
1 1
(17)
0
0
L C
V =
2 2
f
f
2
1
E x =
( )
0
0
0
1
Figure 2 shows the block diagram of the sliding mode
controller, in which the dq0-axis voltage references are
obtained from (7).
Lf Cf
0
L + 3L C
f
n
f
C1
S1
S3
S5
Series Transformer
Lf
and v1, v2 and v3 are new control inputs.
+
_
Vdc
The sliding surfaces with the errors of the indirect
component voltages are expressed as [11]:
C2
Cf
S4
S6
Ln
S2
abc
dq0
S1,2,3,4,5,6
abc
dq0
abc
dq0
s1 = 1 12 e dt
1
i
SVPWM
if0
fq ifd
vc0
vcd
vcq
i
q i
d i0
(12)
Sliding surface
s2 = 2 + 212 + 22 e2 dt
s1
s1
vcq
vc0
v*dvr,d
s2
s3
vcd
s2
s3
u1
s3 = e3 + 313 + 32 e3 dt
-
X
+
dq0
abc
u2
u3
Eq.
(11)
v*dvr,q
-
Eq. (6)
vcd
vcq
i
X
+
if0
vc0
fq ifd
- v*dvr,0
Eq.
(13)
e1 = y1* − y1 e2 = y2* − y2
y3*
e3 = y3* − y3 y1*
X
+
where
,
and
;
,
,
u1eq
u2eq
u3eq
iq
id
i0
y*
y1
y
y3
Eq.
(12)
2 and
are the reference values of the
,
2 and
respectively, and k11, k12, k21, k22, k31 and k32 are the positive
constant gains.
Figure 2. Block diagram of the proposed controller.
By using a sliding mode control theory, the equivalent
control input can be derived as the continuous control input
The system output response to its command is evaluated by
the resonant peak and bandwidth values in the Bode plot.
In order to compare with conventional method, the PI
control technique is also proposed as shown in Figure 3.
Then, the closed-loop transfer function of the cascade PI
controllers is derived as:
s
that
yields.
2
1
1
1
u1eq = Lf Cf v +
ifq
+
+2
v
+
id
iq
1
cd
q
Cf
Lf Cf
Cf
Cf
(13)
k k s2 + k k s + k k s + k k
2
1
1
1
(
)
vdvr
vd*vr
pv
p
pv
i
iv
p
iv i
u2eq = Lf Cf v +
ifd
+
+2
v
+
=
2
cq
d
L C s4 + k C s3 + k C + k k s2 + k k s + k k s + k k
Cf
Lf Cf
Cf
Cf
(
)
(
)
f
f
p
f
i
f
pv
p
pv
i
iv
p
iv
i
1
1
+
(18)
u3eq = Lf Cf v3 +
vc0
i
Cf
L + 3L
C
f
f 0
(
)
f
n
Voltage
Current
controller
controller
To drive the state variables to the sliding surface
+
+
vd*vr
vdvr
+
+
1
Lf s
1
X
X
X
X
X
+
Cf s
s = s = s = 0
s 0, s 0, s 0
, the
1 2 3
+
+
, in the case of
1
2
3
control laws are defined as:
u1 = u1eq + sign s
( )
Figure 3. Control block diagram of DVR using PI control
for voltage and current controllers.
1
1
u2 = u2eq + sign s
(14)
(
)
)
2
2
The Bode plot of the closed-loop transfer function of two
controllers is analyzed in Figure 4. At the low-frequency
range, the two controllers have a unity gain and zero phase
delay. However, The sliding mode control has a lower
resonant peak and a wider bandwidth which results in a
lower overshoot and a faster settling time at the stepwise
load change. Thus, the performance of the sliding mode
control is better than that of the PI control.
u3 = u3eq + sign s
(
3
3
where 1>0, 2>0, 3>0.
The reaching law can be derived by substituting (14) into
(12), which gives
s =− =−
( )
1
( )
1
(15)
In order to determine the stability and robustness,
Lyapunov’s functions which are presented in [12], are
defined as follows:
1
V = s12
1
2
1
(16)
V2 = s22
2
By taking time derivative of V1 and V2, to prove stability,
the following condition must be satisfied
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CONTROL OF DYNAMIC VOLTAGE RESTORER UNDER VOLTAGE SAG AND NONLINEAR LOAD
Under the same simulation conditions of grid voltage
sags and linear loads, as shown in Figure 4(a), the control
performance of the DVR with the proposed method is
shown in Figure 5. Figure 5(c) shows the load voltages,
which are kept at nominal values even though the grid
voltages drop, as shown in Figure 5(a). The output voltages
of the DVR to compensate for the voltage sags are shown
Sliding mode
control
PI control
in
Figure
5(b).
It
is
illustrated in Figure 5(d)–(f) that, the actual values of the
dq0 axis DVR voltage components with the proposed
strategy track their references well, which are much better
than those of the conventional ones, especially with the
method based on the classical PI controllers as shown in
Figure 4 (d) –(f), respectively. In comparison with the PI
controller, the total harmonic distortion (THD) analysis for
load voltage is shown in Table 3, in which the proposed
controller gives better results with lower THD.
PM = 135o
PM = 46o
Figure 4. Bode plot of the closed-loop sliding mode control
and PI voltage controller.
Table 3. Total harmonic distortion (THD) analysis of three-
phase load voltages using PI and proposed controllers.
IV.SIMULATION RESULTS
PSIM simulations have been carried out for the
unbalanced and nonlinear loads to verify the feasibility of
the proposed method. A DC-link voltage at the input of
inverter is 400[V], the switching frequency of inverter is
10[kHz]. The grid voltage is 180Vrms/60Hz. The
parameters of loads and controllers are shown in the Table
1 and Table 2, respectively.
THD (%)
Controller
Type
Linear load
Nonlinear load
Phase Phase Phase Phase Phase Phase
A
B
C
A
B
A
PI control
2.53
2.14
2.69
3.30
2.48
2.80
Proposed
control
1.96
1.83
2.34
2.13
1.96
2.39
Table 1. Parameters of loads
Type of load
Parameters
L = 3 [mH], C = 1000 [F],
R = 30 [Ω]
The performance of the DVR with the conventional PI
control under the conditions of grid voltage sags and
nonlinear loads is shown in Figure 6, in which voltages of
phases a, b, and c also drop to 50%, 75% and 50%,
respectively for 40 [ms] . The DVR output voltage is shown
in Figure 6(b). The waveform of the load voltage is
distorted due to the influence of the nonlinear load Figure
6(c). This shows that the conventional control method do
not respond well. The actual values of the dq0 axis DVR
voltage components are shown from Figure 6(d) to (f),
respectively. The load currents are illustrated in Figure
6(g). On the contrary, for the proposed control method, the
control performance of the DVR is shown in Figure 7. As
can be seen from Figure 7(d) to (f) that, the actual values of
the dq0 axis DVR voltage components with the proposed
strategy follow their references well, which are much better
than those of the conventional ones, as shown in Figure 6
(d) –(f), respectively. Figure 7(c) shows the load voltages,
which are kept at nominal values even though the grid
voltages drop, and no distortion due to the influence of
nonlinear load as shown in Figure 7(a). The output voltages
of the DVR to compensate for the voltage sags are shown
in Figure 7(b).
Nonlinear load
Table 2. Parameters of controllers
Controller Type
Gains of controller
Nonlinear load
kp = 17.5
ki = 13100
kpv = 0.31
Current
controller
Voltage
PI
control
controller
kiv = 892
k11=k21 = k31= 4.4 x103, k12
= k22=k32 = 8.4 x106
Proposed control
The simulation results for the PI control and proposed
control method under the conditions of grid voltage sags
and linear loads are shown in Figures 4 and 5, respectively.
The grid fault is assumed to be unbalanced voltage sags, in
which voltages of phases a, b, and c drop to 50%, 75% and
50%, respectively for 40 [ms].
When the DVR is activated, the DVR output voltages
are injected and load voltages should be kept unchanged.
Moreover, the load voltages after the sag must be sinusoidal
and balanced, like those before pre-sag.
Based on THD analysis results in Table 3 for the case
of using nonlinear loads, it can be seen that THD of the
proposed controller has better results than the PI controller.
Finally, with the same condition, the DVR control in the
proposed method works satisfactorily, since the d-q
component voltages of the DVR are well regulated.
Figure 4 shown the performance of the DVR with the
conventional PI control under the conditions of grid voltage
sags and linear loads. The DVR output voltage is shown in
Figure 4(b) and the load voltage is sinusoidal but still has
some ripple, as shown in Figure 4(c). It is illustrated from
Figure 4 (d) to (f) that, the actual values of the dq0 axis
DVR
voltage components track their references. The load
currents are illustrated in Figure 4(g).
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(a) Grid voltages [V]
(a) Grid voltages [V]
esc
esc
esa
esb
esa
esb
(b) DVR output voltages [V]
(b) DVR output voltages [V]
vcc
vca
vcb
vcc
vca
vcb
(c) Load voltages [V]
(c) Load voltages [V]
vla
vlc
vlb
vla
vlc
vlb
(d) d-axis voltages of DVR [V]
(d) d-axis voltages of DVR [V]
vcd
vcd
vd*vr,d
vd*vr,d
(e) q-axis voltages of DVR [V]
(e) q-axis voltages of DVR [V]
vcq
vcq
vd*vr,q
vd*vr,q
(f) Zero-sequence voltages of DVR [V]
(f) Zero-sequence voltages of DVR [V]
vc0
vc0
vd*vr,0
(g) Load currents [A]
vd*vr,0
(g) Load currents [A]
ila
ilc
ilb
ila
ilc
ilb
Figure 4. Dynamic response of PI control method under the
conditions of grid voltage sags and linear loads. (a) Grid
voltages. (b) DVR output voltages. (c) Load voltages. (d)
d-axis
Figure 5. Dynamic response of proposed control method
under the conditions of grid voltage sags and linear loads.
(a) Grid voltages. (b) DVR output voltages. (c) Load
voltages.
(d)
d-axis
voltages of DVR. (e) q-axis voltages of DVR.
(f)
voltages of DVR. (e) q-axis voltages of DVR.
(f)
Zero-sequence voltages of DVR. (g) Load currents.
Zero-sequence voltages of DVR. (g) Load currents.
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CONTROL OF DYNAMIC VOLTAGE RESTORER UNDER VOLTAGE SAG AND NONLINEAR LOAD
(a) Grid voltages [V]
(a) Grid voltages [V]
esc
esc
esa
esb
esa
esb
(b) DVR output voltages [V]
(b) DVR output voltages [V]
vcc
vcc
vca
vcb
vca
vcb
(c) Load voltages [V]
(c) Load voltages [V]
vlc
vlb
vla
vlc
vlb
vla
(d) d-axis voltages of DVR [V]
(d) d-axis voltages of DVR [V]
vcd
vcd
vd*vr,d
vd*vr,d
(e) q-axis voltages of DVR [V]
(e) q-axis voltages of DVR [V]
vcq
vcq
vd*vr,q
vd*vr,q
(f) Zero-sequence voltages of DVR [V]
(f) Zero-sequence voltages of DVR [V]
vc0
vc0
vd*vr,0
vd*vr,0
(g) Load currents [A]
(g) Load currents [A]
ila
ilc
ilb
ila
ilc
ilb
Figure 6. Dynamic response of PI control method under the
conditions of grid voltage sags and nonlinear loads. (a) Grid
voltages. (b) DVR output voltages. (c) Load voltages. (d)
d-axis
Figure 7. Dynamic response of proposed control method
under the conditions of grid voltage sags and nonlinear
loads. (a) Grid voltages. (b) DVR output voltages. (c) Load
voltages.
(d)
d-axis
voltages of DVR. (e) q-axis voltages of DVR.
(f)
voltages of DVR. (e) q-axis voltages of DVR.
(f)
Zero-sequence voltages of DVR. (g) Load currents.
Zero-sequence voltages of DVR. (g) Load currents.
V. CONCLUSION
In this paper, an advanced control strategy for the DVR
was proposed. The effectiveness of the proposed control
strategy was verified through simulation tests, in which the
load voltage is almost sinusoidal and in-phase with the
supply voltage even under the conditions of grid voltage
sags and linear or nonlinear loads. The feasibility of the
proposed control is verified by simulation results, which
show the better performance than conventional PI method.
For the further work, the experiment must be implemented
with using DSP F28379D to show effectiveness of the
proposed control in the real system.
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Nguyen Trong Huan, Ho Nhut Minh, Van Tan Luong
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[8] Lee S., Chae Y., Cho J., Choe G., Mok H., and Jang D., A
new control strategy for instantaneous voltage compensator
using 3-phase PWM inverter, in Proc. IEEE PESC'98, pp.
248-254, 1998.
Từ khóa - Bộ lưu trữ điện áp động, tải phi tuyến, điều
khiển trượt, sụt áp.
Nguyen Trong Huan was born in
VietNam in 1986. He received his
undergraduate degree in 2010, major in
Electrical and Electronics Technology
from University of Technical Education
of Ho Chi Minh City. In 2014, he
received
the
Master
of
Telecommunication
Engineering
Degree from Posts and Telecommunications Institute of
Technology, Ho Chi Minh City Campus. He is working
at Department of Electrical and Electronic Engineering,
Posts and Telecommunications Institute of Technology,
Ho Chi Minh City Campus, VietNam.
Ho Nhut Minh was born in
Vietnam in 1987. He received his
undergraduate degree in 2010,
major
in
Electronics
&
Telecommunications Engineering
from University of Technical
Education of Ho Chi Minh City. In
2014, he received the Master of
Telecommunication Engineering Degree from Posts and
Telecommunications Institute of Technology, Ho Chi
Minh City Campus. He is working at Department of
Electrical and Electronic Engineering, Posts and
Telecommunications Institute of Technology, Ho Chi
Minh City Campus, VietNam.
[9] Trinh Q.-N. and Lee H.-H., Improvement of unified power
quality conditioner performance with enhanced resonant
control strategy, IET Generation Transmission Distribution,
Vol. 8, No.12, pp. 2114-2123, 2014.
[10] Kim D.-E. and Lee D.-C., Feedback linearization control of
three-phase UPS inverter system, IEEE Transactions on
Industrial Electronics, Vol. 57, No. 3, pp. 963-968, 2010.
[11] Khalifa Al H., Thanh Hai N., Naji Al S., “An improved
control strategy of 3P4W DVR systems under unbalanced
and distorted voltage conditions”, Electrical Power and
Energy Systems, Vol. 98, pp. 233–242, 2018.
[12] Van T. L., Nguyen T. H., Ho N. M., Doan X. N., and
Nguyen T. H., “Voltage Compensation Scheme for DFIG
Wind Turbine System to Enhance Low Voltage Ride-
Through Capability”, 10th International Conference on
Power Electronics (ECCE Asia), pp.1334-1338, 2019.
Van Tan Luong was born in
Vietnam. He received the B.Sc. and
M.Sc. degrees in electrical
engineering from Ho Chi Minh City
University of Technology, Ho Chi
Minh city, Vietnam, in 2003 and
2005, respectively, and Ph.D.
degree in electrical engineering
[13] Slotine J.-J. E. and Li W., Applied Nonlinear Control.
Englewood Cliffs, NJ: Prentice-Hall, pp. 207–271, 1991.
from
Yeungnam
University,
Gyeongsan, South Korea in 2013. Currently, he is
working at Department of Electrical and Electronics
Engineering, Ho Chi Minh city University of Food
Industry. His research interests include power
converters, machine drives, wind power generation,
power quality and power system.
CHIẾN LƯỢC ĐIỀU KHIỂN BỘ LƯU TRỮ ĐIỆN
ÁP ĐỘNG TRONG ĐIỀU KIỆN SỤT ĐIỆN ÁP
LƯỚI VÀ TẢI PHI TUYẾN
Tóm tắt - Trong bài báo này, mô hình điều khiển phi
tuyến cho bộ lưu trữ điện áp động (DVR) được đề xuất để
giảm nhiễu điện áp cho tải dưới điều kiện sụt điện áp lưới
và tải phi tuyến. Đầu tiên, mô hình phi tuyến của hệ thống
bao gồm bộ lọc LC được biểu diễn trong hệ quy chiếu đồng
bộ dq0. Sau đó, quá trình thiết kế bộ điều khiển được thực
hiện bằng cách sử dụng bộ điều khiển trượt, trong đó điện
áp tải được duy trì gần như hình sin bằng cách điều khiển
các thành phần trục dq0 của điện áp ngõ ra bộ DVR. Với
mô hình này, chất lượng điện năng được cải thiện đáng kể
so với bộ điều khiển tích phân tỷ lệ (PI) thông thường trong
SOÁ 04B (CS.01) 2020
TAÏP CHÍ KHOA HOÏC COÂNG NGHEÄ THOÂNG TIN VAØ TRUYEÀN THOÂNG
10
7 trang
19/04/2022
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